Spread spectrum communications system

ABSTRACT

A spread spectrum communication system for direct sequence transmission of digital information having a modulation format which is particularly suitable for indoor communication within residential, office and industrial structures. The modulation format combines BPSK or MSK spreading with FM carrier modulation by data bits and a carrier frequency shift whose magnitude is related to both a chip rate and a spreading sequence length. The carrier, chip clock and data clock are all synchronous and the sequence length is an integral submultiple of the bit length. The system reduces the frequency error between the transmitter chip clock and the receiver chip clock to permit the elimination of a code phase tracking loop in the receiver to reduce the receiver complexity. The receiver has an extended dynamic range which makes possible the reception of very strong signal without an automatic gain control loop (AGC) as well as reducing the time needed for code phase acquisition. The transmission system is highly resistant to CW jamming and short distance multipath effects.

This is a continuation of copending application(s) Ser. No. 07/589,323filed on Sep. 27, 1990 (now U.S. Pat. No. 5,121,407 issued Jun. 9,1992).

The present invention relates to a method and apparatus for thetransmission of digital data in a spread spectrum communications system.In particular, it relates to a spread spectrum communication system foruse in transmitting alarm data within a building or other structurewhich can contain sources of electromagnetic interference and in whichthere is a potential for short distance multipath cancellation effects.

Techniques for achieving direct sequence spread spectrum modulation arewell known The most commonly accepted method involves generating aperiodical, relatively high frequency, repetitive pseudo-noise code (PNcode) and effectively mixing the data signal with this PN code usingexclusive OR gates or a balanced mixer. The resulting signal ischaracterized by very wide bandwidth and very low spectral density. Thedirect sequence spread spectrum modulation is particularly attractivefor data transmission within a building (compared to narrow band radiofrequency carriers) in that the low spectral density characteristic ofsuch signals reduces the tendency for interference with other radiosensitive equipment. Also, spread spectrum techniques have the intrinsicability to reduce interference between multiply reflected versions ofthe transmitted signal, since reflections which differ in phase onlyslightly from the initial signal being decoded result in low signalcorrelation and consequent suppression of such signals. In a similarmanner, the same spread spectrum techniques can be used to suppressother jamming signals which may be present in the medium and interferewith the received signal.

It is further well known that communications within buildings or otherstructures can cause substantial attenuation of the original signal.Consequently, a receiver for such systems must have a wide dynamic rangein order to be able to acquire weak signals as well as strong signals,particularly if the communication is to be performed at random between aplurality of transmitters and receivers.

Both the jamming and multipath phenomena for direct sequence spreadspectrum techniques have been discussed in papers and textbooks andpatent documents. However, most of these have restricted the scope oftheir attention to cases where the jamming signals are introduceddeliberately into the medium and intelligently controlled to cause themost harm. This results in the objective being to determine the methodmost immune to the interference under worst case conditions. It is hasalso usually been tacitly assumed that the data mixing with the PN codeand modulating the radio frequency carrier should result in a modulatedwaveform which will not permit demodulation of the data without priorsynchronization and correlation to the same PN code used for theoriginal modulation. This results in some restriction being placed onthe way the data, PN code and carrier are combined to convey theinformation, as well as restrictions being placed on the relationshipbetween the data rate and the sequence repetition rate.

The multipath delay is usually considered to be relatively long. Thatis, the delay between the original data and the delayed replica isassumed to be much longer than one chip time. The mechanisms devised todeal with this multipath problem are different than those needed tosuppress so called "selective fading" caused by electromagnetic fieldcancellation which occurs when the original signal and the delayedreplica arrive at the receiver in opposite phase. This occurs in thecase where the delay is approximately (2*k+1)*wvl/2, where k is aninteger and wvl is the carrier wavelength where both the original andthe replica have approximately the same amplitude. However, it has beensuggested in the scientific literature that short delays are one of themost important factors to consider for indoor communications.

The above-delineated restrictions are a natural consequence of the majororiginal applications of spread spectrum communications, that is,military applications in generally hostile communications environment inwhich jammers are intelligent and active and communication is attemptedover distances much larger than building interiors. Furthermore, inthese cases, a major concern is a need to hide or at least provide a lowprobability of detection of the attempted communication by enemyeavesdroppers.

Only recently has the Federal Communications Commission (FCC) permittedthe utilization of spread spectrum communications techniques forcommercial applications. The FCC has now permitted such operations inthree separate frequency bands. Thus, there is a need for a spreadspectrum communications system which addresses the needs of commercialapplications and which are not restricted by the needs of militaryapplications.

SUMMARY OF THE INVENTION

It is the general object of the present invention to provide an improvedmethod and apparatus for transmitting spread spectrum signals.

Another object of the invention is to provide a method and apparatus fortransmitting spread spectrum signals which has jamming rejectionsubstantially better than expected for a given spreading sequencelength.

A further object of the invention is to provide a method and apparatusfor transmitting spread spectrum signals in which short distancemultipath cancellation of the type encountered in residential and officebuildings is eliminated or substantially reduced.

Yet another object of the present invention is to provide a method andapparatus for transmitting spread spectrum signals in which there is areduction of the frequency error between the transmitter chip clock andthe receiver chip clock which permits the elimination of a code phasetracking loop in the receiver, which in turn substantially reducesreceiver complexity.

A further object of the present invention is to provide a method andapparatus for transmission of spread spectrum signals in which there isan extension of the dynamic range of the receiver which makes possiblethe reception of very strong signals without an automatic gain controlloop (AGC) while at the same time reducing the time needed for codephase acquisition.

Another object of the present invention is to provide a method andapparatus for transmitting spread spectrum signals that provide thereceiver with immunity from interference from CW signals generated bypersonal computer crystal oscillators or the like.

Another object of the invention is to provide a method and apparatus fortransmitting spread spectrum signals in which data, chip clock andcarrier are synchronous.

A further object of the present invention is to provide a method andapparatus for transmitting spread spectrum signals in which the bitperiod length (Tb) is N (where N is an integer greater than 1) timeslonger than the pseudo noise (PN) sequence period (Ts) which results ina modified spectrum of the spread signal and consequently the spectrumof the jammer being spread by the receiver's despreading mixture.

Yet another object of the present invention is to provide a method andapparatus for transmitting spread spectrum signals in which thetransmitter center frequency is frequency modulated at a data rate witha low deviation and in addition is frequency shifted at a lower rate bya large increment relative to the chip rate and sequence length and inwhich the same data packet consisting of a preamble and data is repeatedafter each frequency shift.

A further object of the present invention is to provide a method andapparatus for transmitting spread spectrum signals in which the datapacket consists of a preamble and data following the preamble, thepreamble being longer than the data and comprising a period in which thecarrier is not modulated and the data comprising a period in which databits modulate the carrier center frequency.

A still further object of the present invention is to provide a methodand apparatus for transmitting spread spectrum signals in which themodulated carrier is then spread to the desired transmission bandwidthby applying a PN code sequence to a balanced mixer or other device toobtain BPSK spreading.

Yet another object of the invention is to provide a receiver forreceiving a spread spectrum signal in which the resultant IF spectralcomponents of a CW jamming waveform present in the RF section of thereceiver are separated in frequency by an amount larger than thebandwidth occupied by the desired despread signal.

A still further object of the present invention is to provide a methodand apparatus for transmitting a spread spectrum signal in which asingle CW jammer can produce at most only one spectral component of thespread jammer present in the passband of the narrow band FM receiverfollowing the despreading mixer.

These and other objects, advantages and features are achieved by amethod of transmitting digital data. A high frequency carrier isgenerated at a carrier frequency. The frequency of said carrier ismodulated by data bits assembled in data packets of predeterminedformat. Modulating the frequency of said second carrier frequency bydata bits assembled in data packets of said predetermined format. Thecarrier is spread during a transmission time, by a predetermined PNsequence having a period Ts related to a bit time Tb, where Tb=N * Ts, Nbeing an integer>1. This generates a spread carrier spectrum comprisingmany spectral components separated by 1/Ts in which the amplitudes ofsaid components are reduced by the spreading function. Components andthe modulation imposed upon them are separable from other components bya narrow band filter having a bandwidth less than 1/(2*Ts).

Another aspect of the invention includes a transmitter for transmittingdigital data. A carrier frequency generator means generates a carrierfrequency signal. A pseudo random sequence generator means generates apredetermined PN sequence. A carrier spread means spreads power ofcarrier frequency by said pseudo noise sequence where the spread carriergenerates spectral components separated by a frequency interval greaterthan 1/Ts where Ts is a pseudo noise (PN) sequence period.

A further aspect of the invention includes a receiver for a spreadspectrum signal. A means for receiving a spread carrier radio signalgenerates a spread carrier electrical signal. A means generates apredetermined pseudo noise sequence (PN sequence)combined with a localoscillator signal. A multiplying means coupled to said receiving meansand to said generating means multiplies said spread carrier electricalsignal by said predetermined PN sequence for collapsing the bandwidth ofthe received spread carrier when the local PN sequence phase is inagreement with the sequence phase imposed on the received spread carrierby a transmitter means. This spreads any jamming signals which arereceived along with the transmitted spread signal into many componentsseparated by 1/Ts intervals and reduces their amplitude by the spreadingfunction. A narrow band FM receiver means having a bandwidth less than1/(2*Ts) coupled to an output of said multiplying means recovers data.

A still further aspect of the invention includes a system fortransmitting and receiving digital data. A means generates a highfrequency carrier. A means modulates the frequency of said carrier bydata bits assembled in data packets of predetermined format. A meansspreads said carrier during a transmission time, by a predetermined PNsequence having a period Ts related to a bit time Tb where Tb=N * Ts, Nbeing an integer>1 and Ts being a pseudo noise (PN) sequence period.This generates a spread carrier sepctrum comprising many spectralcomponents separated by 1/Ts. The amplitudes of the components arereduced by the spreading function. A means receives the transmitteddata. A means multiplies said transmitted spread carrier in a receiverby a locally generated predetermined PN sequence and a local oscillatorfrequency for collapsing the bandwidth of the received spread carrierwhen the local PN sequence phase is in agreement with the sequence phaseimposed on the received spread carrier.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of a transmitter in accordance with thepresent invention;

FIGS. 2a-2e are timing diagrams for signals generated within thetransmitter of FIG. 1, FIG. 2a illustrates the power control signal,FIG. 2b illustrates the data signal, FIG. 2c illustrates the dividercontrol signal, FIG. 2d illustrates the crystal oscillator frequency andFIG. 2e illustrates the carrier frequency;

FIGS. 3a-3d are timing diagrams for the data and pseudo-noise (PN)sequence signals of the transmitter of FIG. 1, FIG. 3a illustrates thePN sequences, FIG. 3b illustrates the clock pulses from the PN sequencegenerator, FIG. 3c illustrates the data clock and FIG. 3d illustratesthe data bits;

FIGS. 4a-4d illustrate the spectral envelopes and close-in spectrasignals transmitted by the transmitter of FIG. 1, FIG. 4a illustratesthe transmitted envelope at time T1, FIG. 4b illustrates the transmittedenvelope at time T2, FIG. 4c illustrates the close-in spectra at time T1and FIG. 4d illustrates the close-in spectra at time T2;

FIG. 5 is a schematic diagram of a receiver for the signals transmittedby the transmitter of FIG. 1;

FIGS. 6a-6c are timing diagrams for signals generated within thereceiver of FIG. 5; FIG. 6a illustrates the frequency shift signal, FIG.6b illustrates the frequency of the reference oscillator and FIG. 6cillustrates the frequency of the local oscillator;

FIGS. 7a-7f illustrate the spectra of the received signals of thereceiver of FIG. 5, FIG. 7a illustrates the spectrum of the signalreceived at the antenna at time T1, FIG. 7b illustrates the spectrum ofthe signal received at the antenna at time T2, FIG. 7c illustrates theenvelope of the spread jamming signal at time T1, FIG. 7d illustratesthe envelope of the spread jamming signal at time T2, FIG. 7eillustrates the close-in spectrum of FIG. 7c and FIG. 7f illustrates theclose-in spectrum of FIG. 7d;

FIGS. 8a-8e illustrate the short multipath effect in the spread spectrumsystem, FIG. 8a illustrates a PN sequence, FIG. 8b illustrates a spreadcarrier, FIG. 8c illustrates a delayed spread carrier; FIG. 8dillustrates the despread sum of the carriers of FIG. 8b and 8c, and FIG.8e illustrates the spectrum of the signal of FIG. 8d;

FIGS. 9a-9d show an illustrative example of a system designed to complywith the FCC modulation mask, FIG. 9a illustrates the spectrum at timeT1, FIG. 9b illustrates the spectrum at time T2, FIG. 9c represents theclose-in spectra of FIG. 9a, and FIG. 9d represents the close-in spectraof FIG. 9b.

DETAILED DESCRIPTION

A communication system according to the present invention is composed ofa transmitter and a receiver. Referring to FIG. 1, the transmitter isgenerally shown as 100. The high frequency carrier signal is generatedin the transmitter 100 by a voltage controlled oscillator (VCO) 110. Thevoltage controlled oscillator 110 is part of a phase lock loop (PLL) 113which also comprises divider 122, phase detector 120, loop filter 118and reference oscillator 124. The output of VCO 110 is coupled to theinput of a divider 122 via lines 111 and 119. The output of the divideris coupled to phase detector 120 via line 121. The output of referenceoscillator 124 is coupled via line 123 to second input of phase detector120. The output of the phase detector is coupled through loop filter 118to the control input of VCO 110. A crystal 126 is coupled to referenceoscillator 124 for stabilizing the output thereof. A varactor diode 128is coupled to reference oscillator 124 via line 142.

The circuitry above described in connection with phase lock loop 113 iswell known to those skilled in the art and not need be described indetail. The operation of the circuitry to produce a narrow band FMmodulated carrier is also well known and need not be discussed in detailhere. The carrier signal produced by the phase lock loop 113 is shown inFIG. 2e and will be described below in connection with that figure.

In one embodiment the phase detector 120 may be a 2 input exclusive ORgate having lines 121 and 123 coupled to the inputs and an outputcoupled through loop filter 118 to the control input of VCO 110. Divider122 may be a dual modulus prescaler integrated circuit such as a FujitsuMB501. Such dual modulus prescalers provide two selectabledivide-by-ratios and are commonly utilized as part of the divider in aPLL circuit in order to preserve the channel spacing. Typically the dualmodulus pescaler precedes fully programmable divider circuits to producea divider capable of operating at VHF and UHF frequencies. The divider122 of the present invention can comprise only the dual modulusprescaler to provide a considerable cost saving.

The output of the phase lock loop on line 111 is coupled to multiplier112 which receives a pseudo noise sequence (PN sequence) signal on line147 from PN sequence generator 132. Multiplier 112 serves as a spreadspectrum modulator for modulating the narrow band FM modulated carriergenerated by PLL 113 to produce a spread spectrum signal on line 117.The spread spectrum signal on line 117 supplies the input of radiofrequency amplifier 114 which generates an amplified signal. Theamplified signal is transmitted to the receiver shown in FIG. 5 viaantenna 116. The antenna 116 is coupled to the output of the radiofrequency amplifier 114 via line 115. The clock utilized to generate thePN sequence is derived from the reference oscillator 124 via divider131. Divider 131 is coupled to reference oscillator 124 by line 148 anddivides the frequency of the frequency oscillator by a predeterminednumber P. The output of divider 131 is coupled to the PN sequencegenerator 132 via line 137. Data is applied to control unit 134 on aplurality of lines illustrated as lines 144 and 145. Each of the lines144, 145 is utilized for a different type of input. For example, in aburglar alarm system for a home, input 145 might represent an alarmcondition from an external sensor (not shown). Such external sensor maybe a normally open or normally closed switch or an electronic detectorapplying a voltage on line 145 to indicate an alarm condition. Thesesensors are well known in the alarm art. Therefore, further discussionis omitted. Line 144 might be utilized to indicate tampering with thetransmitter 100. A well known technique is to enclose a normally closedtamper switch in the transmitter housing. Opening the housing opens theswitch contacts to activate the transmitter to transmit a tamper signal.Alternately, it is known to use a normally open switch as a tamperswitch within the transmitter housing which is closed by opening thehousing to transmit a tamper signal. Other possible input signals to thecontrol circuit 134 includes a low battery indicator which would beinternal to the transmitter and indicate that the battery for thetransmitter requires replacement. A signal would be transmitted to acentral unit to advise the operator to replace the battery. The controlunit 134 converts the input data into packets according to apredetermined format, which is then output on line 139 in synchronismwith a data clock. Data packets in the predetermined format are storedin the control unit 134 for each possible message type. The receipt ofone of the possible input signals triggers the control unit 134 togenerate the appropriate data packet. The data clock is generated by PNsequence generator 132 via divider 133. The input of divider 133 iscoupled to the output of PN sequence generator via line 149. The outputof divider 133 is coupled to the data clock input of control unit 134via line 135. In a preferred embodiment, the PN sequence generatorproduces a cyclic signal which is a pseudo noise sequence (PN sequence)of period Ts which is supplied to mixer 112 via line 147. In addition,it produces clock pulses on line 135 at the beginning of each sequence.The data clock signal is produced by dividing the clock pulses by amodulo N counter 133 which is then coupled to control unit 134 via line135. This results in N sequences being generated during one bit and thebits being synchronous with the sequence period. In a preferredembodiment, N is equal to 4 so that 4 sequences are produced during onebit. The data clock and its relationship to the clock pulses and the PNsequence period are shown in FIG. 3 and discussed below. The datapackets on line 139 are shaped by data shaping filter 136 to reducesharp transitions in the digital bit stream which are unsuitable formodulating a carrier. The signal at the output of filter 136 is shown inFIGS. 2b and discussed below in connection with that figure. The outputof filter 136 is then applied via line 140 to the input of summationamplifier 130. A divider control signal 138 is generated by control unit134 on line 138 which controls the operation of divider 122 to divide bya number M or by M+1. This signal is also applied to the second input ofsummation amplifier 130. The output of summation amplifier on line 141is the signal Umod which is applied to the varactor diode 128 in orderto modulate the narrow band FM carrier generated by PLL 113 in a knownmanner.

Control unit 134 also generates a power control signal on line 146. Inone state, the power control signal disconnects the power from most ofthe transmitter circuitry to conserve power and to assure that thecarrier is transmitted only during two predetermined time periods T1 andT2. This reduces the potential for interference when a plurality oftransmitters are used in a system. Accordingly, the power control signalis applied to a means (not shown), such as an electronic switch, inorder to control the application of power to the transmitter circuitry.

The operation of transmitter 100 will now be described in connectionwith FIGS. 2a through 2d. The control unit 134 is responsive to an inputsignal on lines 144 or 145 to produce a power control signal on line146. The power control signal determines whether the transmitter is inthe ON or OFF state. The signal 202 shown in FIG. 2a has ON periods T1and T2 at which time the transmitter is in the ON state. Between the ONstates T1 and T2 is a short duration OFF state Tg during which time thetransmitter is off. After the transmission during period T2, thetransmitter is again rendered in the OFF state until the nexttransmission is to occur. This signal 202 is applied to the electronicswitch described above to apply power to the remaining portions of thetransmitter during the ON periods. It should be noted that the signalsshown in FIGS. 2b-2e are undefined when the transmitter is in the OFFstate following T2 or preceding T1.

The time Tg between periods T1 and T2 can be set to zero or in apreferred embodiment it can be a randomly assigned value, which is muchlarger than T1 or T2. This reduces the probability of losing a packet ofdata caused by the simultaneous transmission of more than onetransmitter, which may occur in an alarm system where a plurality oftransmitters are utilized and such transmitters transmit at random.

The preferred format of the data signal is shown in FIG. 2b. The datasignal consists of a CW period during which time the signal is set toits nominal value and a data period during which the signal variesaccording to the data bits of the message. The data bits are preceded bypreamble bits and followed by cyclic redundancy check (CRC) bits formessage validation. Notice that the edges of the data bit transitions onsignal 204 are rounded as shown at 206 and 208 by the data filter 136 inorder to be more appropriate for modulating the transmitter withoutcreating many harmonic frequencies. The generation of these harmonicswould be detrimental in a system where many transmitters are utilized.

The frequency of the reference oscillator 124 during the CW period isthe nominal frequency Fr. During the data period, the frequency of theoscillator is deviated in accordance with the data bits to plus andminus a frequency frdev as shown for signal 212 at 214 and 216 in FIG.2d. After the data packet has been transmitted a first time, the controlunit 134 generates a divider control signal 210 on line 138. This signalis coupled to divider 122 by line 143. The signal 210 is utilized tochange the divisor of divider 122 from M to M+1. In addition, the signalis coupled via line 138 to the second input of summation amplifier 130where it is summed along with the output of data shaping filter 136 online 140. The addition of the signal 210 in amplifier 130 changes theoperating point of varactor diode 128. As shown in FIG. 2d, thefrequency of reference oscillator 124 on line 123 is altered by thedivider control signal 210 on line 138 to shift the nominal referencefrequency by df as shown at 212. Changing the divisor of the divider 122and changing the operating point of the varactor diode to change thefrequency of the reference oscillator 124 combine to produce a carrierfrequency which is shown in FIG. 2e. As can be seen from FIG. 2e, thedata for the first data packet has a carrier frequency of MFr whereasthe data for the second data packet has a frequency of (M+1) (Fr+df) orFr+ (M+1)*df higher than the frequency of the carrier for the first datapacket. As stated previously, the transmission of the data twice on twoseparate frequencies differing by odd multiples of N/2*Ts causes no CWjamming signal spectral components being present in the receiver passband to corrupt the message on at least one of the two frequencies toguarantee that the message will be properly received. This requires thatthe loop bandwidth of the PLL is wide enough to accommodate thefrequency shifts of the reference oscillator resulting from modulationby the Umod signal.

FIG. 3 illustrates the timing between the clock pulses on line 137, thedata clock pulses on line 135 and the data bits on line 139 and the PNsequences. FIG. 3a shows a series of identical PN sequences 302, 304,306, 308, 310 and 312. As shown, the period of each PN sequence isdesignated Ts. FIG. 3b shows the PN generation clock pulses on line 137and FIG. 3c shows the data clock pulses on line 135. The clock pulses online 135 are the clock pulses on line 137 divided by N in divider 133.As illustrated, N equal 4. FIG. 3d illustrates the data bits on line139.

As illustrated, there are multiple PN sequences per data bit. Thoseskilled in the art recognize that a single PN sequence or other multiplemay be utilized.

The signal 218 is applied to mixer 112 in FIG. 1 where it is spread bymeans of the PN sequence generated by PN sequence generator 132. Theoutput of mixer 112 is the spread spectrum signal on line 117 which isamplified by RF amplifier 114 and then coupled to antenna 116 by line115.

Referring to FIG. 4, the resulting spectrum of the modulated and spreadsignal, which is time dependent, is shown for appropriately chosen dfand M. The power envelope for the spectrum of the first data packet (T1)is shown in FIG. 4a. The envelope shows the typical (sin x/x)² reductionin peak power from first lobe 410 to second lobe 412 to third lobe 416.The power envelope for the spectrum of the second data packet (T2) isshown in FIG. 4b. The envelope also shows the typical (sin x/x)²reduction in peak power from first lobe 424 to second lobe 422 to thirdlobe 420. The close-in spectra of the modulated signal for the firstdata period T1 is shown in FIG. 4c and the close-in spectra for thesecond data packet during time T2 is shown in FIG. 4d. The lines 426indicate a specific point in the spectra shown in FIGS. 4a and 4b thatare expanded in FIGS. 4c and 4d, respectively.

In FIG. 4c the spectral lines resulting from the spreading of thecarrier at time T1 by a PN sequence having a period Ts and being FMmodulated by the data signal are shown as 430, 432, 434, 436, 438 and440. In FIG. 4d the spectral lines resulting from the spreading of thecarrier at time T2 by a PN sequence having the period Ts and being FMmodulated by the data signal are shown as 460, 462, 464, 466, 468 and470. Comparing FIGS. 4c and 4d it can be seen that the spectral linesare disjoint, that is, they are offset one from the other by (M+1)df.The M+1 factor is the result of changing the divisor of divider 122(FIG. 1). The df factor is the result of changing the operating point ofthe varactor diode 128 by applying the divider control signal on line138 to the second input of summation amplifier 130 (FIG. 1).

This frequency, offset by one-half the spectral line frequency interval,in the example shown, helps in rejecting interfering signals. Offsets ofother than one-half the spectral line frequency interval can be chosen.Assuming a CW interference signal, the interference signal will besubjected to the same PN sequence in the receiver as the desired signaland will thus be converted into a spread spectrum signal. One of thespectral lines produced by this created spread spectrum signal might bewithin the receiver passband and interfere with the reception of thedesired signal. By choosing df appropriately, the total spectrum isshifted so that the spectral line placement during time T2 isapproximately centered between the lines of the spectrum occupied duringtime T1. This guarantees interference free reception during time T1 orT2. This is further explained below in connection with FIGS. 7 and 9.

It should be noted that, for the embodiment shown in FIG. 1, the divider131 has a ratio P equal to 2, and therefore Fr is equal to 2*Fc, whereFc is the chip rate of PN sequence generator. As is well known to thoseskilled in the art, in spread spectrum systems, each bit time issubdivided into a plurality of "chips" which determine the bandwidth ofthe communications channel. The greater the number of "chips" per bit,the greater the bandwidth utilized but the greater the amplitude of theinterfering signal has to be to be effective. That is, the amount ofimmunity from jamming is proportional to the sequence length and theratio between the chip time (1/chip rate) and the bit time (Tb).

Referring to FIG. 5, a receiver 500 designed to take advantage of thespectral characteristics of the above-described transmitted signal isshown. The receiver comprises an antenna 548 coupled to a RF amplifier550 by line 502. The output of RF amplifier 550 is coupled via line 504to a mixer 552 the output of which is coupled via line 506 to a narrowband FM receiver 554. Narrow band FM receiver comprises a band passfilter 553 receiving the output of mixer 552 via line 506. The output ofband pass filter 553 is coupled via line 555 to a first limiter 557. Theoutput of the first limiter 557 is coupled via line 559 to a second bandpass filter 564 the output of which is coupled via line 565 to a secondlimiter 567. The output of the first and second limiters are alsocoupled via line 563 and 575, respectively, to a receiver signalstrength indicator amplifier 569. The output of the second limiter online 571 is coupled to a frequency discriminator 573. The output of thefrequency discriminator on line 528 is coupled to amplifier 582 theoutput of which is coupled via line 530 to integrate and dump filter576. The output of receiver signal strength indicator amplifier 569 iscoupled via line 532 to amplifier 580. The output of amplifier 580 online 534 is coupled to integrate and dump filter 578. The outputs ofintegrate and dump filters 576 and 578 are coupled via lines 536 and 538to A/D converters 574 and 572, respectively. The outputs of A/Dconverters 574 and 572 are coupled respectively by lines 540 and 542 tomicroprocessor 570. Microprocessor 570 generates a signal on line 543 toactivate the automatic frequency control (AFC) by closing switch 569 toconnect line 526 to line 522. Line 522 is an input to summationamplifier 568. The microprocessor also produces a frequency shift signalon line 524 which is a second input to summation amplifier 568. Theoutput of summation amplifier 568 on line 520 is applied to varactordiode 566 in order to control the frequency of reference oscillator 562.The nominal frequency of reference oscillator 562 is controlled bycrystal 564. The reference oscillator outputs a frequency of Fr or Fr+dfon lines 545 and 578. Line 545 is an input to local oscillatorsynthesizer 558 and line 578 is an input to divider 561. Divider 561divides the frequency of the reference oscillator by a value P, which isidentical to that utilized in the transmitter. The output of divider 561on line 516 is the PN sequence generator clock which is input into thereceiver PN sequence generator 560. Receiver PN sequence generator 560also has an input on line 546 which are signals generated bymicroprocessor 570 to control the phase of the PN sequence generator.The local oscillator synthesizer 558 is coupled via line 512 to themicroprocessor 570 to receive synthesizer control signals. The output ofthe local oscillator synthesizer at a frequency Flo is applied via line510 to the input of multiplier (mixer) 556. A second input to multiplier556 is the output of the receiver PN sequence generator 560 on line 514.The output of multiplier 556 is a spread local oscillator signal on line508 which is applied as an input to mixer 552. Microprocessor 570 iscoupled via interface lines 544 to displays and controls for the system(not shown).

The operation of the receiver will now be described in connection withFIG. 5. The signal received from the antenna 548 is coupled via line 502to RF amplifier 550 in which it is amplified and applied to mixer 552via line 504. This signal is multiplied by the spread local oscillatorsignal containing the locally generated PN sequence in multiplier 552.The PN sequence utilized is identical to the one utilized by thetransmitter to spread the signal before transmission. When the phase ofthe PN sequence generated by the receiver PN sequence generator 560 isaligned with a phase of the sequence which is modulating the receivedsignal, the received signal collapses to a narrow band FM modulatedsignal (reconstituted carrier centered at the IF frequency) which can beamplified and demodulated by the narrow band FM receiver 554. The phaseof the PN sequence is controlled by the phase control signals generatedby microprocessor 570 and applied to the receiver PN sequence generator560 via line 546. The microprocessor adjusts the phase of the receiverPN sequence generator 560 to obtain the maximum carrier power at theoutput of the limiting amplifiers of the narrow band FM receiver 554.

In a preferred embodiment of the receiver, the microprocessor 570 firstscans all possible code phases and determines the corresponding carrierpowers. The carrier power is determined through the receiver signalstrength indicator which produces a signal indicative of the carrierpower. This signal is coupled via line 532 to amplifier 580. The signalis amplified and applied to integrate and dump filter 578, the output ofwhich is converted by the A/D converter 572 into a digital signal whichis applied via line 542 to an input of microprocessor 570.

The receiver utilizes a sliding correlator as is well known in the art.It is assumed that the first sequence is being transmitted. The power ofthe incoming waveform is determined for one sequence and this value isstored in the microprocessor 570. The microprocessor then changes thephase of the generated PN sequence by one chip and repeats the process.This is repeated for all possible sequence phases (equal to the numberof chips per sequence). The microprocessor then chooses the phasecorresponding to the strongest carrier power. Optionally, themicroprocessor can move the phase ±1/2 chip to maximize the output forthe exact phase.

The microprocessor 570 determines the optimum value of the PN generatorphase in accordance with a the above described algorithm and then setsthe phase of the PN generator via a signal on line 546. At this time,the microprocessor generates a signal on line 543 which closes switch569 to activate the AFC circuit. The AFC circuit adjusts the frequencyof the reference oscillator 562 to obtain a despread signal whosenominal frequency is ideally the IF frequency Fif on line 506. That is,once the PN sequence generators in the receiver and transmitter are inphase, the carrier frequency produced at the output of mixer 552 may notbe exactly aligned with the IF frequency of the narrow band FM receiver.The AFC circuit is utilized to align the receiver frequency with thetransmitted frequency so that the resultant frequency at the mixeroutput is the chosen IF frequency, in a known manner. The AFC circuit isdisabled until the microprocessor recognizes a valid signal.

The frequency of the local oscillator synthesizer 558 and the PNsequence generator clock on line 516 are both derived from the samereference frequency oscillator 562, having a nominal frequency of Fr.Therefore, the clock frequency of the receiver PN sequence generator 560approaches the clock frequency of the transmitter PN sequence generator132 at the end of AFC acquisition process when the signal on line 506approaches the nominal value Fif. If the transmission time T1 or T2 isappropriately chosen to be short enough, the drift of the receiver andtransmitter will be small. This eliminates the need for trackingcircuitry to insure that the phase of the receiver PN sequence generatoris maintained with respect to the phase of the received signal for theduration of the transmission time, which substantially reduces thecomplexity of the receiver without significant impact on the receiver'sperformance.

In order to accommodate the signals at both transmitter frequencies, thefrequency of the local oscillator synthesizer 558 is switchedperiodically between two frequencies F101 which is equal to Ft×1-Fif anda frequency F102 which is equal to a frequency Ft×2-Fif where Ft×1 isequal M*Fr and Ft×2 is equal to (M+1)* (Fr+df). The term df is thetransmitter increment of the reference oscillator 124 in thetransmitter. The processor obtains this value in two steps. First, theprocessor changes the nominal value of the reference oscillator 562 byapplying a frequency shift signal on line 524 input to summationamplifier 568. This changes the voltage on the varactor diode 566 by anamount needed to change the frequency from Fr to Fr+df. Second themicroprocessor 570 controls the frequency of the synthesizer 558 via thesynthesizer control signal on line 512 to obtain the proper value forFlO. The rate of the switching of the transmission periods T1 and T2 arechosen so that the PN code acquisition can be obtained on bothfrequencies under the worst timing conditions. The sliding correlatorrequires enough time to examine each chip phase for each possible chipsequence. This requires a time equal to the square of the number ofchips times the chip time. For example, if we utilized 32 chips persequence, the minimum time required would be (32)² Tc. In a preferredconfiguration, the CW period during time T1 and time T2 should beapproximately 6*L*Tb*Dc where L is the sequence length, Tb is the bittime and Dc is a designer choice constant which is usually 1 or 2. Thereceiver dwell time Td on one frequency has in the preferred embodimenta time of 2*L*Tb*Dc. The receiver frequency shift signal, the frequencyof the reference oscillator and the frequency of the local oscillatorare shown in FIG. 6. FIG. 6a represents the frequency shift signalgenerated by microprocessor 570 and input to summation amplifier 568 vialine 524. The time Td is the receiver dwell time in the search mode,which has been described above. During time T' the reference oscillatoroutput shown in FIG. 6b is at a frequency Fr. During a time T" thefrequency of the reference oscillator is at a frequency Fr+df. Thefrequency of the local oscillator synthesizer is shown in FIG. 6c.During the time T' it is at frequency of Ft×1-Fif and during the periodT" it is at a frequency Ft×2-Fif.

Upon PN code acquisition the signal on line 506 (FIG. 5) consists of thedespread, narrow band FM modulated signal and potentially a spreadjamming signal whose spectral components are spaced by 1/Ts, where Ts isthe spreading sequence period. The exact placement of the spectralcomponents of the spread jamming signal with respect to the desiredsignal is determined by the jamming frequency prior to spreading.

FIGS. 7a and 7b show the spectrum of the signal received at the receiverantenna during time T1 and T2 with a jamming signal present in thespectrum. The signal 702 in FIG. 7a is the signal transmitted duringtime T1. The signal 704 is the jammer at a power level of P_(j) and afrequency of F_(j). The typical (sin x/x)² envelope produced bymodulating rectangular pulses is shown by the attenuation between lobes702, 706 and 708. The (sin x/x)² characteristic causes lobe 706 to be 13dB lower than the desired signal lobe 702. Lobe 708 is lower than lobe706 in accordance with the (sin x/x)² rolloff. In FIG. 7b, 710represents the signal transmitted during time T2. Reference numeral 712represents the jammer at a frequency F_(j) ' and a power P_(j) '.Reference numerals 714 and 716 represent the (sin x/x)² envelope.Similarly to FIG. 7a, lobe 714 is 13 dB lower than lobe 710 and lobe 716is lower than lobe 714 in accordance with the (sin x/x)² rolloff. FIGS.7c and 7d illustrate the envelope of the spread jamming signal afterconversion at the output of mixer 552 (FIG. 5) during time T1 and T2. Asis well known to those skilled in the art, when a CW signal is passedthrough a mixer of a spread spectrum receiver, it is acted on similarlyto the signal passing through multiplier 112 of the transmitter (FIG.1). That is, the signal is spread by the PN sequence. This produces thespectrum shown in FIGS. 7c-7f. In FIG. 7c, reference numeral 720represents the jammer envelope after the mixer 552 during the time T'.Reference numeral 722 represents the peak power of the jammer signal indb as P_(j) '-10LogL and 724 represents the power P_(j). In FIG. 7d, 730represents the jammer envelope after the mixer 552 during the time T".Reference numeral 732 represents the power P_(j) " and reference numeral734 is the power P_(j) -10LogL.

FIGS. 7e and 7f show the close-in spectrum of the spread jamming signalat the output of mixer 552 (FIG. 5) during time T1 and T2 at points 724(P_(j) ') and 732 (P_(j) ") and also the relationship of the jammingsignal component to the despread desired signal and the receiver IFbandwidth. The spectrum shown in FIGS. 7e and 7f is an expansion of thepoints in FIGS. 7c and 7d along line 728. In FIG. 7e, 740 represents thedespread received signal and 742 represents the bandwidth of the IF bandpass filter. Reference numeral 744 represents the jammer at power P_(j)' 746 and 748 represent the close-in spectrum jammer components duringtime T'. In FIG. 7f, reference numeral 750 represents the despreadreceived signal. Reference numeral 752 represents the bandwidth of theIF band pass filter. Reference numeral 754 represents the jammer signalat a power P_(j) " and reference numeral 756 and 758 represent theclose-in spectrum of the jammer components during time T".

The jamming signal envelope, during the time T1, is centered around thefrequency F_(j) -Flo' where Flo' is the local oscillator frequencyduring time T1, which is above the Fif by approximately F_(j) -Ftx1,where F_(j) is the jammer frequency and Ftx1 is the center frequency ofthe signal transmitted at time T1. Similarly, the jamming signalenvelope during the time T2 is centered around F_(j) -Flo" where Flo" isthe local oscillator frequency at time T2, which is below Fif byapproximately Ft×2-F_(j), where Ft×2 is the center frequency of thesignal transmitted at time T2. This is due to the frequency conversionin the mixer 552 based on changing the frequency of the localoscillator. Because of the (sin x/x)² shape of the envelope, the powerlevel at P_(j) " (which is in the second lobe) is approximately 10 dBlower than the power level at P_(j) ', where P_(j) ' and P_(j) " are thejamming signal spectral component power level translated to the IFfrequency during times T1 and T2, respectively.

The drawings illustrate that if the despread signal spectrum is narrowenough so that the receiver bandwidth is less than 1/(2*Ts) and thefrequency difference between Ft×1 and Ft×2 is equal to Fr plusapproximately 1/(2Ts) then at least one reception, that is, Ft×1 orFt×2, is not corrupted by the jamming signal interference. This isbecause a narrow-bandwidth receiver filter is effectively between thespectral lines of the spectrum jamming signal. Furthermore, the jammingspectral lines are down by the spreading factor plus an additionalfactor related to the spread jamming signals (sin x/x)² spectralenvelope. This can be seen in FIGS. 7e and 7f at P_(j) ' which is in thepassband whereas P_(j) " is not.

As a result, the frequency shift technique is effective in suppressingthe non-intentional CW jamming signals, such as those created bycomputers, terminals, and other devices employing crystal based or otherfrequency determining elements as part of clock timing circuits. Somegain is also obtained in the suppression of narrow band as well as wideband modulated jamming signals due to the spreading factor and result inthe (sin x/x)² envelope of the spread jamming signal caused by thecorrelating mixer.

The frequency shift technique described above is also an effective meansof suppressing short delay multipath signals frequently encountered whena high frequency carrier is used to communicate inside of buildings. Thepresent invention provides good results for both long delay and, moreimportantly, for short delay multipath effects in a spread spectrumcommunication system in which the delays are much shorter than the chiptime. Traditional methods are less effective on this type of multipathdelay. FIG. 8 illustrates the short multipath effect on the spreadspectrum signal. FIG. 8a represents a PN sequence having a chip time ofTc. FIG. 8b represents a spread carrier and FIG. 8c represents a delayedspread carrier. FIG. 8d represents the sum of an original and a delayedsignal assuming that the power of the time-delayed signal is comparableto the power of the original signal and that the delay is such that thetwo signals are in opposite phase at the receiver antenna. FIG. 8erepresents the spectrum of the resultant signal of FIG. 8d. If the powerof the spread carrier and the delayed spread carrier is each A, then thepower of the signal in FIG. 8d is 2A as shown at 810. In FIG. 8d thebulk of the signal has been cancelled since the signals only add duringchip transitions and then only for the duration of the time delayT_(Del) as shown in 810, 820, 830, 840 and 850. The resultant power ofthe signal shown in FIG. 8d is much smaller than the power of theoriginal signal. If the power that would be available is P_(TOT), thenthe power available at 830 or 840, for example, is ##EQU1## whereT_(Del) is the delay time between the signals of FIGS. 8b and 8c. In apractical example for a chip rate of 5 million chips per second and atime delay of 20 nanoseconds, the power of the signal shown in FIG. 8dis 5 times smaller than the power of the original signal. In addition,the spectrum of the resultant signal is much wider, as shown in FIG. 8eand therefore only a part of the total power of the signal can passthrough the narrow band receiver filters. In the example discussedabove, only one tenth of the energy originally present is available inthe detector. Thus, the total loss of power, considering theabove-described loss of 80% of the original power, is thereforeapproximately 50 times or 17 dB as shown at 880. The power at this lobeis P_(TOT) '/(T_(c) /T_(Del)).

The frequency shift arrangement described above provides for bettersuppression of the multipath effect because it introduces frequencydiversity which changes the phase relationships of the original andtime-delayed signals. This reduces or eliminates the cancellation effectshown in FIG. 8d. For a carrier frequency of approximately 1 GHz, thetotal loss encountered may be less than 3 dB as opposed to 17 dB in theexample above (FIG. 8e).

Another feature of the receiver is related to the reception of strongsignals above the dynamic range of the receiver signal strengthindicator (RSSI). In the case when the signal is very strong andconsiderably above the noise level, there may be no need to despread thesignal before detection. According to FIG. 4, if the spectrum of eachcomponent of the spread spectrum signal is narrow enough (in practice,less than 1/Ts) the data can be recovered directly from any of thespectral components without prior despreading. This permits turning offthe local PN generator when a predetermined signal strength level hasbeen exceeded for a predetermined period of time thus disabling theacquisition process. Therefore the evaluation of the RSSI signal, whichaids the acquisition, is irrelevant. The signal received is so strongthat peaks of the spectral lines would be received without despreading.Each spectral lines carries the full modulation; thus allowing the datato be extracted from a single spectral line. This known technique hasvery practical implications since the RSSI amplifier usually saturatesat signal levels which are much smaller than the maximum signal level atwhich the limiters and the FM discriminator can be expected to functionproperly. The effective dynamic range of the receiver can be extended inthis manner without the necessity for automatic gain control circuitry.In addition, the acquisition time can be reduced for all transmitterswhich are close enough to the receiver to produce signals at thereceiver antenna which are strong enough to actuate this detectionmechanism. This type of a narrow band signal spectrum associated witheach of the spectral lines can be obtained by using an appropriatelychosen deviation together with a bit time Tb much greater than the PNsequence period Ts.

The signals generated in a preferred embodiment of the system accordingto the present invention are shown in FIG. 9. FIGS. 9A and 9B show thespectral envelope of the transmitted signal and its relationship to theFederal Communications Commission (FCC) modulation mask for spreadspectrum transmission. The modulation mask allots the frequency band of902 to 928 MHz and requires that the signal at the boundaries beapproximately 20 dB below the level of the transmitted frequencies Ft×1and Ft×2. In FIG. 9, the following parameters were chosen:

    M=128

    P=2

    L=32

    Fr=7.12 MHz and

    N=4

These choices result in component spacing of the spread signal of 111.25kHz and a df of 111.25/(2*129) or approximately 430 Hz. The receiverreference oscillator and the transmitter reference oscillatorfrequencies are changed from 7.12 (T1) to 7.12043 MHz (T2) or byapproximately 60 ppm. The spectra in FIGS. 9a and 9b follow the (sinx/x)² rolloff. The lobe 906 at the 902 MHz lower boundary 912 isapproximately 19.5 db down from the lobe 902 at frequency Ft×1=911.36MH_(z) during time T1. The lobe 924 at the 928 MHz upper boundary 926 isapproximately 20.2 dB down from the lobe 920 atFxT2=918.44+129*df=918.496 MHz during time T2. Thus minimal filtering isrequired at the lower boundary and no filtering is required at the upperboundary to meet the FCC modulation mask requirements. Thissignificantly reduces the complexity and cost of the transmitter. Thispermits the use of nonlinear Class C or similar amplifiers in thetransmitter which results in reduced power consumption and thus makesthe transmitter more suitable for battery operation.

FIGS. 9c and 9d show the close-in spectra within lines 930. As can beseen these spectra are separated by 111.25 KHz. Comparing FIG. 9d to 9c,the spectra of FIG. 9d are offset by approximately 1/2 chip=56 KHz. Areceiver bandwidth (BW) of 50 kHz fulfills the requirement that BW isless than 1/(2Ts) and allows for the reception of the modulated signalat a data rate of 111.25/4 kilobits per second or approximately 27.8kilobits per second.

While a particular embodiment of the present invention has beendisclosed herewithin, certain changes and modifications will readilyoccur to those skilled in the art. All such changes and modificationscan be made without departing from the invention as defined by theappended claims.

We claim:
 1. A method of transmitting digital data, comprising the stepsof:a) generating a radio frequency carrier; b) modulating the frequencyof said carrier by data bits assembled in data packets of predeterminedformat; c) spreading said carrier during a transmission time, which apredetermined PN sequence having a period Ts related to a bit timeTbwhere

    Tb=N*Ts,

N being an integer>1 Ts is a pseudo noise (PN) sequence period;whereby aspread carrier sepctrum comprising many spectral components separated by1/Ts is obtained, the amplitudes of said components being reduced bysaid spreading function, said components and the modulated imposed uponsaid components being separable from other components by a narrow bandfilter having a bandwidth less than 1/(2*Ts).
 2. The method according toclaim 1, further comprising the steps of:d) receiving the transmitteddata; and e) multiplying said transmitted spread carrier in a receiverby a locally generated predetermined PN sequence, for collapsing thebandwidth of the received spread carrier when the local PN sequencephase in agreement with the sequence phase imposed on the receivedspread carrier thereby spreading any jamming signals which are receivedalong with the desired transmitted signal into many components separatedby 1/Ts intervals and reduced in amplitude by the spreading function. 3.The method according to claim 1, wherein step c) generates spectralcomponents of the spread carrier that are separated by a frequencyinterval greater than 1/Ts.
 4. The method according to claim 1, whereina chip clock producing a signal at a frequency of 1/Ts and the carrierfrequency are synchronously derived from one frequency source.
 5. Themethod according to claim 4, wherein the data frequency is synchronouslyderived from said one frequency source.
 6. The method according to claim1, wherein said predetermined data packet format comprises a preambleperiod during which said carrier frequency is equal to a nominalfrequency and a data period during which the carrier frequency ismodulated by said data bits.
 7. A transmitter for transmitting digitaldata comprising:a) carrier frequency generator means for generating acarrier frequency signal; b) modulator means for modulating said carrierby data bits; c) pseudo random sequence generator means for generating apredetermined PN sequence; d) carrier spread means for spreading powerof said carrier frequency by said pseudo noise sequence, said spreadcarrier generating spectral components separated by a frequency intervalgreater than 1/Ts where Ts is a pseudo noise (PN) sequence period. 8.The transmitter according to claim 7 wherein Ts is related to the bittime Tb where

    Tb=N*Ts,

N being an integer>1.
 9. The transmitter according to claim 7 furtherincluding a phase lock loop means for deriving said first carrierfrequency from a crystal reference oscillator.
 10. The transmitteraccording to claim 7 wherein said transmitter further includes means forderiving chip frequency of 1/Ts and said carrier frequency from a singlereference frequency source.
 11. The transmitter according to claim 7wherein said transmitter further includes means for deriving chipfrequency of 1/Ts, said first carrier frequency and a data clockfrequency from a single reference frequency source.
 12. The transmitteraccording to claim 7 wherein said transmitter further includes means forgenerating said formated data packet for modulating said carrierfrequency, said data packet comprising a preamble period during whichthe carrier frequency is equal to a nominal value and data period duringwhich the carrier frequency is deviated according to the correspondingdata bit values.
 13. A receiver for a spread spectrum signalcomprising:a) a wide band receiving means for receiving a spread carrierradio signal for generating a spread carrier electric signal; b) meansfor generating a predetermined PN (pseudo noise) sequence; c)multiplying means coupled to said receiving means and said generatingmeans for multiplying said spread carrier electrical signal by saidpredetermined PN sequence, for collapsing a bandwidth of the receivedspread carrier when the local PN sequence phase is in agreement with thesequence phase imposed on the received spread carrier by a transmittermeans, thereby spreading any jamming signals which are received alongwith the transmitted spread signal into many components separated by1/Ts intervals and reduced in amplitude by the spreading function; d)narrow band FM receiver means coupled to an output of said multiplyingmeans for recovering data and having a bandwidth less than 1/(2*Ts)where Ts is a period of said PN sequence.
 14. The receiver according toclaim 13 wherein each of said receiver means further includes means forderiving a chip frequency of 1/Ts and a carrier frequency from a singlereference frequency source.
 15. The receiver according to claim 14wherein each of said receiver means further includes means for derivinga chip frequency of 1/Ts, a carrier frequency, and a data clockfrequency from a single reference frequency source.
 16. The receiveraccording to claim 14 wherein each of said receiver means furtherincludes means for automatic frequency control (AFC) and means foractivating said AFC upon synchronization of a phase of said PN sequencewith a phase of the PN sequence modulating said receiver signal.
 17. Thereceiver according to claim 13 wherein each of said receiver meansfurther includes signal strength indicator means to detect power of saidreceived signal.
 18. A system for transmitting and receiving digitaldata, comprising:a) means for generating a radio frequency carrier; b)means for modulating the frequency of said carrier by data bitsassembled in data packets of predetermined format; c) means forspreading said carrier at said first and said second frequency, during atransmission time, by a predetermined PN sequence having a period Tsrelated to a bit time Tbwhere

    Tb=N*Ts,

N being an integer>1 Ts is a PN (pseudo noise) sequence period;whereby aspread carrier sepctrum comprising many spectral components separated by1/Ts is obtained, the amplitudes of said components being reduced bysaid spreading function; d) means for receiving the transmitted data;and c) for multiplying said transmitted spread carrier in said receivermeans by a locally generated predetermined PN sequence, for collapsingthe bandwidth of the received spread carrier when the local PN sequencephase in agreement with the sequence phase imposed on the receivedspread carrier; f) narrow band FM receiver means coupled to an output ofsaid multiplying means for recovering data and having a bandwidth lessthan 1/(2*Ts).
 19. The system according to claim 18, wherein said meansfor spreading said carrier generates spectral components of the spreadcarrier that are separated by a frequency interval greater than 1/Ts.20. The system according to claim 18, wherein a chip clock producing asignal at a frequency of 1/Ts and the carrier frequency aresynchronously derived from one frequency source.
 21. The systemaccording to claim 20, wherein the data frequency is synchronouslyderived from said one frequency source.
 22. The system according toclaim 18, wherein K is greater than a sequence length for generating afrequency shift substantially equal to or greater than a chip frequencyof 1/Ts.
 23. The system according to claim 18, wherein saidpredetermined data packet format comprises a preamble period duringwhich said carrier frequency is equal to a nominal frequency and a dataperiod during which the carrier frequency is modulated by said databits.